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CC Poon and Edward Chui, Motorola SPS, Hong Kong
Interfacing two systems that operate at two arbitrary voltages is a challenging
problem; there is no guarantee that one side operates at a voltage higher than the other
side. Usually, the interface is an open-collector or open-drain type with just two
transistors connecting back to back (Figure 1a). VX
is the lower of the two operating voltages. If you know which side has the lower operating
voltage, the interface design is straightforward. If either side can have the lower
operating voltage, you have to extract the lower one. Without the use of an op amp, you
can use a diode-based circuit (Figure 1b). The 1N4148 is
good for most applications. If a higher current capability is necessary, you can use the
1N4001. If the lower operating voltage is around 1V, D3 should be a Schottky
diode, such as the 1N5817 or MMBD701, and D1 and D2, can be normal
PN-junction diodes.
If level translation is necessary in one direction, you can use half of the circuit for
open-drain translation, which is equivalent to simple TTL. This simple circuit is fast (Figure 2a). When driving one standard load on a real pc
board, which has approximately 10 to 20 pF of total load capacitance, the rise and fall
times are fast when you view it with a scope. Circuit performance is better than that of
the traditional bipolar inverter, which needs a compen sation capacitor to assist
turn-off.(Replacing the bipolar transistor with an enhancement MOSFET can eliminate the
capacitor but results in long rise and fall times and a longer delay.) The TTL-like
circuit uses only pullup resistors, which may further save pc-board space because you can
use multiple pullup resistors in one resistor pack.
For a logic high-to-low transition, the delay is just the turn-on time of the
transistor. For a low-to-high transition, RC effects don't appear until the output rises
to about 0.5V below the lower supply voltage, VX, when translating up (Figure 2b). Before that, the output tracks the input with
only a VCE(SAT) drop, which is analogous to a cascode amplifier. The effect of
turning off a saturated transistor does not manifest itself except when translating from
below 1V to 5V.
The TTL-like circuit in Figure 2a also works well for
translating from high to low voltages (VB<VA). For a high-to-low
transition, the delay is just the turn-on time of the transistor. You can replace the
pullup resistor by an active transistor to increase driving strength (Figure 2c). You must pay attention to VEBO(BR),
which must not exceed the rated value because a violation results in premature failure of
the transistor. For most small signal transistors, VEBO(BR) is typically 4 to
5V. Therefore, you should take care when down-shifting from 12 to 5V, such as between CMOS
analog circuits and 5V logic.
The switching transistor can be MPS2369A to MPS3646 for high-speed switching. You can
use the 2N3904 or BC547 for low-power applications. A 2N5458 can replace the pullup
resistor at the collector if active pullup is necessary. The best 1-to-5V shifting driver
in the laboratory produces a typical symmetric delay of 6 nsec using three-fourths of an
MPQ2369 when driving a 74AC541 buffer (Figure 3). (DI
#2290)
Anil Kumar Maini and Nita Sen, Defence Science Centre, Delhi, India
The conventional advantages of lasers—coherence, monochromaticity, and extreme
compactness—make laser diodes popular in most of their potential applications.
Biomedical diagnostics and high-resolution-spectroscopy applications exploit laser diodes'
wavelength tunability. These applications use the laser wavelength's sensitivity to drive
current and operating temperature—0.025 nm/ and 0.3 to 0.4 nm/°C for diodes emitting
approximately 700 nm, respectively. However, this sensitivity also puts a stringent
requirement on the stability of these parameters. The resolution with which the output
wavelength can vary depends on the stability or accuracy of the sensitivity parameters.
The drive-current sensitivity of 0.025 nm/mA suggests that a 10-MHz accuracy, which is a
modest requirement, necessitates a drive-current stability of 0.7 µA, which is equivalent
to 7 ppm, assuming a drive current of 100 mA.
The low-cost and small circuit in Figure 1 is a stable
laser-diode driver with an optional modulation-input facility. The circuit features soft
start, soft decay, and immunity to noise transients. The circuit operates from a single
supply of 12V and uses a quad "Norton" op amp, the LM3900. IC1A, Q1,
R1 to R4, and C1 constitute the basic constant-current
source with the magnitude of current depending on the dc voltage present at Point X and
the value of R4. The dc voltage at X stabilizes the drive current in the
feedback mode against any variations in the IV characteristics of the laser diode and the
power-supply voltage. The feedback signal consists of a proportional voltage appearing
across sense resistor R5, which noninverting IC1B amplifies by a
gain 15. The output of this amplifier drives differential amplifier IC1C. One
of the inputs to IC1C is a bandgap-derived reference voltage. The differential
amplifier has a gain of 15.
A small change in the drive current results in a large change in the control voltage at
X in a direction that restores the current to the nominal value. The closed-loop gain of
the circuit is approximately 20. Changing the reference voltage to the differential
amplifier, which is the voltage at Point Y, changes the nominal value of the current.
Although the chosen component values produce a drive current of 60 mA, the circuit can
produce drive current of 50 to 80 mA. R6, C2, and Q2
provide soft-start and soft-decay features. The observed soft-start and soft-decay times
are approximately 200 and 500 msec, respectively. A lowpass filter comprising R7
and C3 has a cutoff of approximately 10 Hz in the feedback loop to provide
immunity to fast transients.
Tests show that the circuit has a stability better than ±0.05%/hour. The observed
short-term current stability is better than ±0.02%. The observed variation in drive
current for a ±2V variation in power-supply voltage is less than 0.1%. Experimental
measurements by connecting an appropriate resistance across R4 introduce a step
change of 2 mA. Measurements also show the resultant change in current and a closed-loop
gain of approximately 20. The Toshiba (www.toshiba.com)
TOLD-9211 laser diode emitting approximately 4 mW tested the circuit for a drive current
of 60 mA at 670 nm. The circuit fits into a DIP-like, eight-pin metal package. (DI #2291)
Jerzy Chrzaszcz, Warsaw University of Technology, Poland
Of the many solid-state voice-storage chips available, the
ChipCorder family from
Integrated Storage Devices (ISD, San Jose, CA) is one of the most user-friendly. A single
chip integrates nonvolatile voice memory, a microphone preamplifier, and an output stage
capable of driving a 16 Ohm loudspeaker. A simple interface allows you to record and play
messages under manual or µP control. The configuration in Figure
1 allows you to copy the contents of one chip to another. For single units, consider
recording each chip anew. For regular production, you could purchase a gang programmer
from ISD. However, for prototyping and short production runs, the circuit in Figure 1 offers an attractive cost/performance ratio.
The programmer accommodates 25xxx-series chips with recording time as fast as 120 sec.
It consists of two ZIF sockets, control logic, and some passive components. The output
signal from the Master chip traverses R1, R2, C1, and C2
to the Target inputs. R3 and C3 couple the Target's preamplifier and
amplifier. R4 and C4 provide an AGC delay to the Target (Consult the
ISD data sheet for details.) Strapping of the control pins ensures that the Master can
only play back and the Target can only record; however, to avoid hazardous transient
states, you should lock the voice chips in their sockets before switching on the power.
The controller is configured as an asynchronous state machine that uses just two 7474
flip-flops. Its simplicity results from the highly autonomous operation of the voice
chips. This design uses the M4 function mode (A4, A8, and A9 pulled high), which provides
sequential addressing of the messages without controller intervention. Closing the Copy
switch starts Master playback simultaneously with Target record (CE set low, LED1 on).
When the Master issues "End of Message" (EOM), recording stops (CE set high,
LED1 off) and the EOM marker automatically goes into the Target's memory. The cycle
repeats whenever you close the Copy switch, so you can copy messages one by one.
After you copy the entire Master contents, the overflow (OVF) line goes low, signaling
an overflow condition. This signal turns LED2 on. Closing the Reset switch clears the
flip-flop and resets the internal address counters of the voice chips. The capacitor
across the Reset switch generates a power-on reset pulse. The circuit uses internal
clocking for the voice chips; therefore, actual message addressing may differ slightly
from copy to copy. This effect is negligible, as long as all messages fit into memory and
your application uses a sequential-access mode. However, even if your target system
directly addresses the chip, you can still use this programmer if the Master copy provides
some spacing between consecutive messages. You thus avoid the potential errors of
overlapping EOM markers past the nominal start address of the next message. Also, because
of AGC regulation, the sound level of Target messages may, and probably will, differ from
the original. (DI #2272).
David Salerno, Unitrode Corp, Merrimack, NH
Lithium-ion batteries are rapidly gaining popularity in portable applications because
of their superior energy density, low self-discharge rate, and high cell voltage. When you
use one Li-ion battery to power a 3.3V dc/dc converter; however, you encounter a problem,
because the battery voltage can be higher or lower than 3.3V. When fully charged, a Li-ion
cell has approximately 4.2V output; when fully discharged, the voltage is approximately
2.5V. Therefore, you cannot use a simple buck or boost topology with a single inductor to
generate a regulated 3.3V output. Some designs boost the voltage to approximately 4.3V and
then use a low-dropout regulator to produce the 3.3V. This approach is inefficient, and
efficiency is a crucial consideration in battery-powered applications. The circuit in Figure 1 offers a solution to the problem.
The circuit works by referencing the positive terminal of the battery to system ground
and using a flyback topology with a single low-cost inductor to generate 3.3V, with
respect to system ground. IC1, a UCC3954, is a fixed-frequency, 200- kHz
voltage-mode PWM converter that includes an internal 0.15V MOSFET switch. Gate drive for
the FET comes from bootstrapping off the 3.3V output. The converter works efficiently over
a load of 0 to 650 mA. Note that the input and output filter capacitors should be low-ESR
tantalums or OSCONs. Output ripple is lower than 1% at maximum load. The inductor value is
not critical; 33 mH is a good compromise between size and efficiency.
The compensation components (R1, C1, and C2) ensure
stability and provide good transient response over a wide load. For applications in which
no sudden changes in load current occur, you can use a simpler, dominant-pole compensation
method. In this case, you can omit R1 and C1 and increase C2
to 0.039 mF. The UCC3954 includes a low-battery-warning output and a shutdown input. The
low-battery warning is a current-limited, open-drain output that turns on when the battery
voltage approaches the shutdown threshold of the IC. You can use it to turn on an LED or
to drive an input to a µP to provide an alert that power will soon be lost.
To enable IC1, you should pull the shutdown input up to output ground. When
this input is left open, it pulls down to the battery () potential, and IC1's
quiescent current reduces to less than 1 mA. To prevent overdischarging the Li-ion
battery, IC1 automatically turns off when the input voltage drops to less than
2.5V, and the quiescent current reduces to 30 mA. Although IC1 is designed for
use with single-cell Li-ion batteries, you could also power the converter using three
nickel-based rechargeables or three alkaline cells in series. As with any high-frequency
converter, layout and grounding critical to proper operation. Keep all connections as
short as possible, and use a ground plane. (DI #2263).
Paul J Rose, Mental Automation Inc, Bellevue, WA
The test circuit in Figure 1 efficiently drives various
capacitive loads, such as memory cells and simple capacitors, so that you can observe
their leakage effects. Essentially, the circuit is a pulsed and variable current source
acting as a charge pump. A pulsed voltage source drives a one-shot oscillator. This
one-shot drives two MOSFET switches that convert the 10V rail-to-rail output of the
oscillator to the desired rail-to-rail voltage drive—in this case, 15V—for the
controlled current mirror with the same voltage- switching polarity. The current mirror
drives the variable load.
R1 and C1 determine the timing pulses that IC1's one-shot
oscillator produces. When IC1's output is high, Q1 is on, and Q2 is
off. The floating drain of Q2 causes the emitter and base of Q3 to
have the same potential, so that Q3 is off. Then, the pnp current mirror of Q4
and Q5 turns on to drive the variable load of R2 and C2
high. R3 controls the charge rate of the load. As you make R3 smaller,
the current mirror provides more current to the load to charge it up faster, as required
for testing.
When the output of the one-shot is low, Q1 is off, and Q2 is on.
In this case, the drain of Q2 is at ground potential, and the base of Q3
is at a lower potential than its emitter so that Q3 turns on. Current through Q3
flows through R3, causing a voltage rise at the bases of Q4 and Q5,
which turns them off. Turning off Q4 and Q5 disconnects the variable
load from its power supply so that the load is free to bleed stored charge through R2.
This pulsing charge pump has three unique features: It can generate various pulse
widths, the variable resistor in the coupled-collector circuit of the pnp current mirror
provides a variable charging rate, and the circuit accommodates separate voltage drives
for the one-shot oscillator and load using MOSFET switches. Any signal-propagation delay
or asymmetrical switching effects through the pump cause no adverse latency effects if the
pulse periods are on the order of 100 msec or more. Note that you can replace Q1
and Q2 with a variable gain buffer follower as Figure
1 indicates. In this case, one 15V supply drives the follower.
Spice simulations, using models developed in-house and by semiconductor vendors,
confirm the circuit's operation (Figure 2). (DI #2248)
Acknowledgment
The author thanks Mental Automation Inc, whose ECAD tools he used to implement and test
the circuit. The company provided the author the company time to submit this idea. The
author also thanks Seattle Silicon Inc (Bellevue, WA) for permitting him to publish this
Design Idea, which the author built and tested in Seattle Silicon's laboratory as part of
a test project.
Boris Khaykin, Candid Logic Inc, Madison Heights, WI
The simple tester in Figure 1 detects short circuits on
assembled pc boards and also rings out cables and harnesses. The short finder has a narrow
zone of threshold uncertainty and very low "insertion" voltage and current, and
it's not confused by capacitors. The circuit uses an LM10, an IC that combines a precision
200-mV reference, a reference buffer, and an independent, high-quality op amp. It can
operate from supply voltages of 1.1 to 40V. The op amp in this design serves as a
comparator. The voltage from the reference buffer, via R2, creates a
positive-going bias shift at the balance input and a negative-going bias shift at the
comparator's inverting input.
When the tested circuit resistance exceeds 2V, the negative-going bias overrides the
positive-going bias, and the comparator delivers 0V to the buzzer. Otherwise, the
comparator delivers full output voltage to the buzzer to indicate a short circuit. R1
limits the current to the circuit under test to less than 1 mA. The circuit's current
drain is less than 300 µA with open test probes and approximately 2 mA with the probes
shorted together. Open-circuit voltage is 200 mV, which is less than the turn-on voltage
for pn junctions. If desired, you can set the voltage as low as 15 mV by adding 18 Ohm
resistance between pins 2 and 3 of IC1. However, the quiescent current
increases to 1 mA.
You can change the resistance threshold by changing the value of R2. With
the values shown, the threshold is approximately 2V. The supply voltage can be within 1.1
to 30V, depending on the buzzer's voltage range. You can use any piezo buzzer with current
consumption lower than 20 mA. You can easily build the short finder as an adapter for a
DMM, provided that the DMM has a continuity function (Figure 2).
Upon detection of a resistance that is less than 2V, the short finder delivers a virtual
negative resistance to the DMM. By nature, this signal is lower than any DMM continuity
threshold (which is always positive); therefore, the circuit works with any DMM. R3
limits the current to the DMM's input circuitry to approximately 1 mA. (DI #2264).
Lyle Williams, Electronic Technical Services, New Orleans, LA
Parallel LC circuits that you tune by changing capacitance have a nonlinear
frequency-versus-voltage or frequency-versus-shaft-position characteristic. The frequency
of an analog-tuned circuit is proportional to the reciprocal of the square root of the
tuning capacitance. When you tune a bandwidth that is say, 5% or less of the center
frequency, the frequency-versus-capacitance over this limited band is essentially linear.
Because the frequency is proportional to capacitance, it's desirable to have a linear
capacitance-versus-shaft-position or capacitance-versus-voltage characteristic. A
mechanical variable capacitor can provide a linear capacitance-versus-rotation
characteristic. However, mechanical tuning capacitors are expensive and large and have
limited reliability.
You frequently use varactor diodes for voltage control of capacitance. But their
capacitance-versus-voltage characteristic is approximately logarithmic, not linear. In the
days of vacuum tubes, designers used reactance-tube circuits for automatic frequency
control in FM receivers and for modulating FM transmitters. It's possible to make the
capacitance of the circuit proportional to the transconductance (gm) of the
tube. Over a certain bias range, the tube's gm is proportional to the grid bias
voltage. You can build such a reactance circuit using FET or bipolar transistors (Figure 1). The current in the drain circuit is in quadrature
with the drain voltage because of the feedback elements R1 and C1.
As a result, the drain circuit emulates a capacitor.
You can control the capacitance using voltage, via potentiometer R2, or by
adjusting R1's shaft position. (You should set the unused potentiometer to
maximum.) The L and C values in this controlled tuned circuit are chosen for a frequency
range in the vicinity of the 49m short-wave band. The controlled LC circuit serves to tune
a regenerative-type receiver. The tuning dial for this radio is linear—a feature
uncommon in analog receivers. A modern version of the regenerative receiver can provide
performance comparable with that of a simple superheterodyne receiver. Regenerative
receivers are unique in that they require only one LC resonant tuning circuit. A superhet
requires at least two resonant circuits that must track each other as you tune the
receiver.
You can change the frequency band of a regenerative receiver by switching a single
two-terminal inductor. You could also use Figure 1's
controlled tuned circuit to tune an RF amplifier, a filter, or an oscillator. The
reactance circuit produces a maximum capacitance of CR=gm3R13C1.
R1 is the total resistance of potentiometer R1. The reactance of C1
should be much larger than R1 at the frequency of interest: XC1>>R1.
Figure 2 shows the result of using voltage tuning via
potentiometer R2. The curve is linear from 0.1 to 1.3V. The change in frequency
that accrues in this voltage range is 6.2 to 6.07 MHz for a 190-kHz bandwidth.
Figure 3 shows the results of shaft tuning. The bend at
the lower end of the curve comes from the potentiometer characteristic. The curve is
linear throughout the entire tuning range, which is 210 kHz wide. In the
reactance-"tube" circuit, it's desirable to use a transistor with high output
impedance. In this respect, a pentode vacuum tube with an output impedance of
approximately 750 kOhm is superior to a transistor. However, MOSFETs have a considerably
higher gm than tubes. The transistor's gm determines the amount of
change in capacitance that is possible. The maximum gm of a 3N200 MOSFET is
15,000 mmho, and the output impedance is 13 kOhm. (DI #2267). |
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