The recent advent of Analog Devices’ ADCMP60x family of comparators has filled a gap between the less-than-1-µsec-response comparators consuming 100 to 200 mW and those exhibiting approximately 1-µsec response, requiring about one-thousandth that power. The ADCMP60x comparators exhibit a low value of the product of propagation-delay-by-supply-current drain; possess rail-to-rail input and output operation; and offer a variety of options for hysteresis, latch-mode operation, and shutdown mode. Some of them also have inherent level-translating capability. Moreover, the ratio of propagation delays for the positive and negative transitions at the output is close to the ideal value of 1 within 8% tolerance for the ADCMP600, ADCMP601, ADCMP602, and ADCMP603 and with in a 6.7% tolerance for the ADCMP608 and ADCMP609 members of the family (Reference 1).

This ratio is important in applications in which both positive- and negative-output-level transitions are equally significant. Figure 1 shows one such circuit. Voltage-level transitions at the output of the detector indicate changing of the sign of the first derivative of the input signal; in other words, the circuit detects time positions of peaks and valleys in the input-voltage waveform. The detector circuit uses an ADCMP601 for IC₂, and IC₁ is an Analog Devices AD8007 current-feedback amplifier. IC₁ connects as a voltage follower with an antiparallel combination of Schottky-barrier switching diodes, D₁ and D₂, between the output and the inverting input of the amplifier. Comparator IC₂’s inputs connect to the source of the input voltage and to the output of the current-feedback amplifier. This configuration enhances the voltage difference of V_IN–V_A between inputs of the comparator. It performs this enhancement in a steplike manner at the instant, or region, at which the sign of slope of the input signal changes. This voltage difference is a measure of the double-forward voltage of diodes D₁ and D₂ at their forward current, which you derive from V_IN/R_F.

You use a current-feedback amplifier as IC₁ because a dynamic current flows into its inverting input even when you connect it as a voltage follower. The values of the R_S and R_F resistors are those that Reference 2 recommends for a gain of 1. You needn’t worry about instability due to the presence of antiparallel diodes in the feedback path of the current-feedback amplifier. These diodes increase the value of feedback resistance to more than 499Ω. Whenever the input voltage is only approximately 0V, the frequency-gain response of IC₁ for an R_F value greater than 499Ω remains flat.

An analysis of the response of the voltage follower in Figure 1 to a harmonic input voltage uses ω/ω_T and ω=2πf, where f is the input-voltage frequency and ω_T is the radial transition frequency of the amplifier. At the radial-transition frequency, the ratio of Z_M (the magnitude of the amplifier’s transimpedance) to R_F drops to one. This simplification leads to an equation for the delay, t_D, in Figure 2:

where V_F is the forward voltage of diode D₁, V_m is the amplitude of input voltage, R_m0 is the dc transresistance of the current-feedback amplifier, and ΔΦ is the electrical-error angle in radians. The period of input harmonic voltage, T in Figure 2, represents 2π radians. The final error of the detector is ΔΦ, which decreases by a factor of . This reduction occurs because the necessary operating overdrive over the midpoint of the steplike transition in the V_A(t) voltage that the comparator requires is more than an order of magnitude less than the value of V_F.