SDRs (software-defined radios) provide enormous flexibility, permitting you to change modes or waveforms at will. This Design Idea focuses on the “exciter” portion of a moderate-bandwidth SDR (Figure 1). The RF carrier or transmitter IF enters the quadrature modulator, and the modulated output exits for further frequency translation or amplification, depending on the details of the design. The DSP section generally works with analytic signals—in this case, signals with real and imaginary parts—at baseband. These signals may have started out as a voice speaking into a microphone that attaches to an ADC, or they may have started out as data from a computer. Regardless of the signals’ origin, the DSP performs calculations on the stream of numbers, performing filtering, perhaps adding signaling tones or packetizing the data, and converting the stream into the final I and Q modulating signals. For moderate bandwidths, a stereo sigma-delta DAC or codec provides the conversion to analog signals and performs some additional filtering on the signal. Such filtering is often necessary because the quadrature modulator comprises a pair of mixers. These mixers translate any noise at baseband frequencies directly to the modulator’s output.

Output noise is problematic. The FCC (Federal Communications Commission) sets spectral masks or adjacent-channel-power-ratio requirements on some services, such as land mobile radio. These requirements govern the allowed spectrum of a transmission and vary according to the bandwidth of the channel and the frequency of transmission. Their function is always the same, however: They limit the interference to other users on nearby channels to the transmitter. Meeting the spectral mask is a regulatory requirement; you cannot certify a radio without proving that it meets this requirement, and, without this certification, you cannot legally sell it. Figure 2 shows a sample spectral mask, 47 CFR 90.210 G, with a normalized X axis to show the offset from the center of the channel and a normalized Y axis to show the unmodulated carrier output. This mask applies to the 800-MHz SMRS (specialized-mobile-radio service) in which channels are 25 kHz apart but signals can occupy only 20 kHz.

The unmodulated carrier first transmits at the center of the mask, and the top of the mask adjusts to correspond with the output power of the transmitter. You then turn on the modulation, thereby spreading the spectrum. The resulting spectrum must fall below the mask line in all places.

A close examination of Figure 2 shows some interesting features. On the carrier trace, the sampling-frequency spurs appear at ±19.2 kHz away from the center. The modulated spectrum is also interesting. The filter in the sigma-delta DAC causes the nearly vertical roll-off at approximately ±10 kHz. The mounds that appear around ±12 kHz and gradually roll off are spectral regrowth, which nonlinearities in the high-power amplifier cause.

Many moderate-bandwidth SDRs need a translator between the sigma-delta DAC’s single-ended output and a typical balanced-input quadrature modulator. It is frequently desirable to follow up the DAC output with a hardware filter that removes the DAC’s high-frequency noise and ensures compliance with spectral-mask requirements. Further complicating things, the optimal common- and differential-mode output voltages of the DAC are likely to differ from those that the modulator requires. An easy scaling factor does not relate common- and differential-mode voltages.

Handling all of these considerations with a conventional approach can require as many as four operational amplifiers with multiple filter sections per I or Q channel. The filters require close component matching to guarantee that carrier and single-sideband suppression—key measures of quadrature-modulator ideality—do not degrade as a function of baseband frequency. The Linear Technology LTC1992, on the other hand, addresses the problem in a single section. Linear shows a fully balanced approach to the problem in its data sheet (Reference 1).

It turns out, however, that a fully balanced approach is unnecessary. The circuit in Figure 3 has excellent phase and amplitude balance between the output channels and eliminates some critical component-matching requirements. Pin 2 is set for the desired common-mode output voltage, and the DAC’s midpoint voltage connects through an input resistor to Pin 8. Note that any mismatch between the input voltage and the midpoint voltage appears at the outputs and causes asymmetrical swing. This application bypasses Pin 7. The filter is a passive single-pole circuit cascaded with an inverting Sallen-Key filter, but other topologies are feasible.

Figure 4 shows the measured frequency response of the positive channel with respect to ground. The apparent 6-dB loss is a result of looking at only half the differential-output voltage; when you examine the full balanced output, the net gain is 0 dB. Figure 5 shows the measured deviation from an ideal equal-amplitude, 180° phase shift between the positive and the negative outputs. The agreement in the critical 300-Hz to 3-kHz range is less than 0.1 dB and 0.1°. Even at 50 kHz, the error is less than 0.5 dB and 1°.

The authors gratefully acknowledge the assembly assistance of Deb Girard.