Second-generation current conveyors feature wide signal bandwidth, linearity, wide dynamic range, simple circuitry, and low power consumption. Hence, designers employ several implementations of current mode in these devices for realizing various functions. A previous Design Idea introduced a second-generation dual-output current-controlled conveyor to create oscillators (Reference 1). Unfortunately, these circuits aren’t available as ICs, but you build them from discrete components. Figure 1 illustrates an active building block of such a circuit, which the following equations characterize: I_Y=0, V_X=V_Y+I_XR_X, I_Z+=I_X, and I_Z–=–I_X. You can express the parasitic resistance at terminal X as R_X=V_T/2I_B, where V_T is the thermal voltage and I_B is the bias current of the conveyor that is tunable over several decades. Figure 2 shows the bipolar implementation of the circuit.

The circuit provides an extra degree of freedom in the sense that the control over the frequency of oscillation can be through both current and voltage. The circuit in the previous Design Idea provides various advantages, it this new circuit not only retains all those essential advantages, it also provides an extra feature of voltage controllability of frequency of oscillation. Additionally, you can control the condition of oscillation using the conveyors’ bias currents.

Figure 3 shows the proposed sinusoid-oscillator circuit. You can obtain the characteristic equation for the circuits as follows: S₂C₁C₂R_X1R_X2+SC₂R_X2–SC₂R_X1+K=0, where K is the voltage multiplier. Satisfying Barkhausen’s criteria—that the loop gain is unity or greater and that the feedback signal arriving back at the input is phase-shifted 360°—the required condition for oscillation is R_X1=R_X2, and the frequency of oscillation is

Clearly, you can use the gain buffer to vary the frequency of oscillation, which is the area in which this circuit differs from the earlier Design Idea. You can use both current and voltage to control the voltage multiplier. The circuit lets you vary the voltage multiplier by adjusting bias currents I_B3 or I_B4 (Figure 4). For voltage control over K, you can use another circuit simply by using a noninverting op amp and replacing the resistors with MOSFETs working in that triode region. That approach simulates voltage-controlled resistors.

The circuit in Figure 2 underwent testing with a PR100N PNP transistor and an NPN NP100N transistor of the bipolar arrays ALA400 and a dc supply of ±3V (Reference 2).

The circuit requires only two current-controlled conveyors, two grounded capacitors, and a voltage multiplier; it requires no floating capacitors and no external resistors, which makes the circuit’s power consumption lower than that of RC oscillators. For a conventional bipolar-transconductance operational amplifier, the transconductance, g_m, is I_B/2V_T. Comparing this figure with the equivalent value of I_B, the transconductance of the bipolar-transconductance op amp is four times less than that of a dual-output current-controlled conveyor. Thus, the power consumption of the current-controlled-conveyor-based circuit is about four times less per active device than that of the op-amp-based circuit. The sensitivity study shows that S^ωC _{K;RX1;RX2;C1;C2}=–½; ωc sensitivities are hence less than unity, which is an attractive feature of this circuit. Remember that creating an accurate oscillator model requires modeling equations to be nonlinear, and meeting the Barkhausen criteria is a necessary condition for oscillation. Oscillator circuits may latch up and never oscillate even if you satisfy the Barkhausen criteria.