Design a 100A active load to test power supplies

-September 22, 2011

Design a 100A active load to test power supplies imageYou use an active-load test circuit to ensure that a power supply for a microprocessor or for other digital loads supplies 100A transient currents. This active load can provide a dc load for a power supply, and it can rapidly switch between dc levels. These transient loads simulate the fast logic switching in the microprocessor.

Ideally, your regulator output is invariant during a load transient. In practice, however, you will encounter some variations, which become problematic if allowable operating-voltage tolerances are exceeded. You can base your active-load circuit on previous designs of wideband loads that operate at lower currents (Reference 1). This approach allows you to design a closed-loop, 500-kHz-bandwidth, 100A active load having linear response.

Conventional active-load circuits have shortcomings (Figure 1). The regulator under test drives dc and switched resistive loads. Monitor the switched current and the output voltage so that you can compare the stable output voltage versus the load current under both static and dynamic conditions. The switched current is either on or off. You cannot control it in the linear region as it changes.

Design a 100A active load to test power supplies figure 1

You can further develop the concept by including an electronic-load switch control (Figure 2). The input pulse switches the FET through a drive stage, generating a transient load current from the regulator and its output capacitors. The size, composition, and location of these capacitors have a profound effect on transient response. Although the electronic control facilitates high-speed switching, the architecture cannot emulate loads that are between the minimum and the maximum currents. Additionally, you are not controlling the FET’s switching speed because doing so introduces wideband harmonics into the measurement that may corrupt the oscilloscope display.

Design a 100A active load to test power supplies figures 2-3

Transient generator

Placing Q1 within a feedback loop allows true, linear control of the load tester (Figure 3). You can now linearly control Q1’s gate voltage, allowing you to set an instantaneous transient current at any point and to simulate nearly any load profile. Feedback from Q1’s source to control amplifier A1 closes a control loop around Q1, stabilizing its operating point. The instantaneous input-control voltage and the value of the current-sense resistor set Q1’s current over a wide bandwidth. You use the dc-load-set potentiometer to bias A1 to the conduction threshold of Q1. Small variations in A1’s output result in large current changes in Q1, meaning that A1 need not supply large output excursions. The fundamental speed limitation is the small-signal bandwidth of the amplifier. As long as the input signal stays within this bandwidth, Q1’s current waveform is identical in shape to A1’s input control voltage, allowing linear control of the load current. This versatile capability permits you to simulate a wide variety of loads.

You can improve this circuit by adding some components (Figure 4). A gate-drive stage isolates the control amplifier from Q1’s gate capacitance to maintain the amplifier’s phase margin and provide low delay and linear current gain. A gain-of-10 differential amplifier provides high-resolution sensing across the 1-mΩ current-shunt resistor. You can design a power-dissipation limiter that acts on the averaged input value and Q1’s temperature. It shuts down the FET’s gate drive to preclude excessive heating and subsequent destruction. Capacitors can be added to the main amplifier to tailor the bandwidth and optimize the loop response.

Design a 100A active load to test power supplies figure 4

You can develop a detailed schematic based on these concepts (Figure 5). The main amplifier, A1, responds to dc and pulse inputs. You also send it a feedback signal from A3 that represents load current. A1 sets Q1’s conductivity through the Q4/Q5 gate-drive stage, which is actively biased using A2. The voltage drop across the gate drive’s input diodes would be high enough to fully turn on Q4 and Q5. To prevent this overdrive, reduce the voltage across the lower diode with Q3. Amplifier A2 determines the gate-drive-stage bias by comparing Q5’s averaged collector current with a reference and controlling Q3’s conduction, thus closing a loop. That loop keeps the voltage drop across the bases of Q4 and Q5 to a value well under 1.2V, and servos that value until Q4 and Q5 have a 10-mA average collector-bias current.

Design a 100A active load to test power supplies figure 5
Click to enlarge.

 Click here to read more of Jim Williams' contributions to EDN.
The duty cycle of the load overheats if it is on for too long. You can fashion a protection circuit with techniques that high-power-pulse-generator designers use (references 2, 3, and 4). Feed comparator IC1 the average input-voltage value. It compares that voltage to a reference voltage set with the dissipation-limit-adjust potentiometer. If the input duty cycle exceeds this limit, comparator IC1 turns off the FET gate drive through Q2. Thermal switch S1 provides further protection. If Q1’s heat sink gets too hot, S1 opens and disconnects the gate-drive signal. By diverting Q4’s bias voltage, transistor Q6 and the zener diode prevent Q1 from turning on if the −15V supply is not present. A 1-kΩ resistor on A1’s positive input prevents amplifier damage should you lose the 15V power supply.

Design a 100A active load to test power supplies figure 6Trimming optimizes the dynamic response, determines the loop’s dc baseline idle current, sets the dissipation limit, and controls the gate drive’s stage bias. The dc trims are self-explanatory. The loop-compensation and FET-response ac trims at A1 are subtler. Adjust them for the best compromise between loop stability, edge rate, and pulse purity. You can use A1’s loop-compensation trimming capacitor to set the roll-off for maximum bandwidth and accommodate the phase shift that Q1’s gate capacitance and A3 introduce. The FET-response adjustment partially compensates Q1’s inherent nonlinear-gain characteristic, improving the front and rear pulses’ corner fidelity (see sidebar “Trimming procedure”).

Circuit testing

You initially test the circuit using a fixture equipped with massive, low-loss, wideband bypassing (Figure 6). It is important to do an exceptionally low-inductance layout in the high-current path. Every attempt must be made to minimize inductance in the 100A path. You should get good results after you properly trim the circuit if you minimize inductance in the high current path (Figure 7). The 100A-amplitude, high-speed waveform is pure, with barely discernible top-front and bottom-rear corner infidelities (see sidebars “Verifying current measurement” and “Instrumentation considerations”).

Design a 100A active load to test power supplies figures 7-12

Design a 100A active load to test power supplies figure 13To study the effects of ac trim on the waveform, you must perform deliberate misadjustments. An overdamped response is typical of excess A1 feedback capacitance (Figure 8). The current pulse is well-controlled, but the edge rate is slow. Inadequate feedback capacitance from A1 decreases the transition time but promotes instability (Figure 9). Further reducing the trim capacitance causes loop oscillation because the loop’s phase shift causes a significant phase lag in the feedback. Scope photos of uncontrolled 100A loop oscillation are unavailable. The event is too thrilling to document. Overdoing the FET’s response compensation causes peaking in the corners of the waveform (Figure 10). Restoring the ac trims to nominal values causes a 650-nsec rise time, equivalent to a 540-kHz bandwidth, on the leading edge (Figure 11). Examining the trailing edge under the same conditions reveals a somewhat-faster 500-nsec fall time (Figure 12).

Design a 100A active load to test power supplies figures 14-15

Design a 100A active load to test power supplies figures 16-18Layout effects

If parasitic inductance is present in the high-current path, your design cannot remotely approach the previous responses. You can deliberately place a tiny, 20-nH parasitic inductance in Q1’s drain path (Figure 13), which will cause an enormous waveshape degradation deriving from the inductance and the loop’s subsequent response (Figure 14a). A monstrous error dominates the leading edge before recovery occurs at the middle of the pulse’s top. Additional aberration is evident in the falling edge’s turn-off. The figure’s horizontal scale is five times slower than the optimized response (Figure 14b). The lesson is clear: High-speed 100A excursions do not tolerate inductance.

Regulator testing

After you address the compensation and layout issues, you can test your power-supply regulator (Figure 15). The six-phase, 120A LinearTechnology Corp LTC1675A buck regulator acts as a demonstration board. The test circuit generates the 100A load pulse (Trace A of Figure 16). The regulator maintains a well-controlled response on both edges (Trace B of Figure 16). The active load’s true linear response and high bandwidth permit wide-ranging load-waveform characteristics. Although the step-load pulse in Figure 16 is the commonly desired test, you can generate any load profile. A burst of 100A, 100-kHz sine waves is an example (Figure 17). The response is crisp, with no untoward dynamics despite the high speed and current. You could form a load even from an 80-μsec burst of 100A p-p noise (Figure 18). The load circuit has high accuracy, compliance, and regulation specifications (Figure 19 and Table 1).
Design a 100A active load to test power supplies figure 19

Design a 100A active load to test power supplies table 1

  1. Williams, Jim, “Load TransientResponse Testing for Voltage Regulators,” Application Note 104, Linear Technology Corp, October 2006.
  2. “Overload Adjust,” HP-214A Pulse Generator Operating and Service Manual, Hewlett-Packard, figures 5 through 13.
  3. “Overload Relay Adjust,” HP-214A Pulse Generator Operating and Service Manual, Hewlett-Packard, 1964, pg 5.
  4. “Overload Detection/Overload Switch,” HP-214B Pulse Generator Operating and Service Manual, Hewlett-Packard, March 1980, pg 8.

Author's biography
Jim Williams was a staff scientist at Linear Technology Corp, where he specialized in analog-circuit and instrumentation design. He served in similar capacities at National Semiconductor, Arthur D Little, and the Instrumentation Laboratory at the Massachusetts Institute of Technology (Cambridge, MA). He enjoyed sports cars, art, collecting antique scientific instruments, sculpture, and restoring old Tektronix oscilloscopes. A long-time EDN contributor, Williams died in June 2011 after a stroke.
Trimming procedure

Trimming Figure 5’s circuit is a seven-step procedure that must be performed in order. Out-of-sequence adjustments are permissible if you have adjusted the dissipation-limiter circuitry.

  1. Set all adjustments to midrange except A1’s feedback capacitor, which should be at full capacity.
  2. Apply no input. Bias Q1’s drain from a 1V-dc supply. Turn on the power and trim the baseline current for 0.5A through Q1. Monitor this current with an ammeter in Q1’s drain line.
  3. Turn off the power. Lift Q2’s source lead and let it float, which disables the dissipation-limit circuitry, leaving Q1 vulnerable to damage from inappropriate inputs. Follow the remainder of this step in strict accordance with the instructions. Turn on the power, bias Q1’s drain from a 1V supply, apply a −0.1V-dc input, and monitor Q1’s drain current with an ammeter. Trim the gain adjustment for a 10.5A meter reading. The trimming gain at only 10% of scale mandates the tight trim targets. This limitation is undesirable but less painful than trimming at 100% of scale, which would force astronomical—and brief—dissipation in Q1 and the 1-mΩ shunt resistor. Make this adjustment fairly quickly because Q1 dissipates 10W. Turn off the power and reconnect Q2’s source lead. It is worth mentioning that the primary uncertainty necessitating gain trimming is the sense line’s mechanical placement at the 1-mΩ shunt resistor.
  4. Turn on the power with no input and with Q1’s drain unbiased. Trim the IQ adjustment for 10 mV at A2’s positive input measured with respect to the −15V rail. Turn off the power.
  5. Bias and bypass Q1’s drain in accordance with Figure 6. Set the drain’s dc-power supply for 1.5V output and turn on the power. Apply a 1-kHz, −1V-amplitude, 5-μsec-wide pulse. Slowly increase the pulse width until IC1 trips, shutting down circuit output and illuminating the power-limit LED. Tripping should occur at approximately a 12- to 15-μsec pulse width. If it does not, adjust the dissipation-limit potentiometer to bring the trip point within these limits. This step sets the allowable full-amplitude 100A duty cycle at approximately 1.5%.
  6. Under the same operating conditions as those in Step 5, set the input pulse width at 10 μsec and adjust A1’s capacitive trim for the fastest positive-going edge obtainable at A3’s output without introducing pulse distortion. Pulse clarity should approach that in Figure 7 with somewhat degraded top-front and bottom-rear corner rounding.
  7. Adjust the FET-response compensation to correct the corner rounding in Step 6. Some interaction may occur with Step 6’s adjustment. Repeat steps 6 and 7 until A3’s output waveform looks like that in Figure 7.

Verifying current measurement

Theoretically, Q1’s source and drain current are equal. Realistically, they can differ due to the effects
of residual inductances and the 28,000-pF gate capacitance. A3’s indicated instantaneous current
could be erroneous if these or other terms come into play. You can verify that the source and the drain currents are equivalent (Figure A). Add a top-side, 1-mΩ shunt and a gain-of-10 differential amplifier to duplicate the circuit’s bottom-side current-sensing section. The results should eliminate concern over Q1’s dynamic-current differences (Figure B). The two 100A pulse outputs are identical in amplitude and shape, promoting confidence in the circuit’s operation.

Instrumentation considerations

The pulse-edge rates in the main article are not particularly fast, but high-fidelity response requires some diligence. In particular, the input pulse must be cleanly defined and devoid of parasitics, which
would distort the circuit’s output-pulse shape. A1’s 2.1-MHz input RC (resistance/capacitance) network filters the pulse generator’s preshoot, rise-time, and pulse-transition aberrations, which are
well out of band. These terms are not of concern. Almost all general-purpose pulse generators should perform well.

A potential offender is excessive tailing after transitions. Meaningful dynamic testing requires a rectangular pulse shape, flat on the top and the bottom within 1 to 2%. The circuit’s input band-shaping filter removes the aforementioned high speed-transition-related errors but does not eliminate lengthy tailing in the pulse flats. You should check the pulse generator for this issue with a
well-compensated probe at the circuit input. The oscilloscope should register the desired flat-top- and flat-bottom-waveform characteristics. In making this measurement, if high speed-transition-related events are bothersome, you can move the probe to the band-limiting 300-pF capacitor. This practice is defensible because the waveform at this point determines A1’s input-signal bandwidth.

Some pulse-generator output stages produce a low-level dc offset when their output is nominally at its 0V state. The active-load circuit processes such dc potentials as legitimate signals, resulting in a
dc-load baseline-current shift. The active load’s input scale factor of 1V=100A means that a 10-mV zerostate error produces 1A of dc baseline-current shift. A simple way to check a pulse generator for this error is to place it in external-trigger mode and read its output with a DVM (digital voltmeter). If offset is present, you can account for it by nullifying it with the circuit’s baseline-current trim. You could also use a different pulse generator.

Keep in mind parasitic effects due to probe grounding and instrument interconnection. At pulsed
100A levels, you can easily induce parasitic current into “grounds” and interconnections, distorting displayed waveforms. Use coaxially grounded probes, particularly at A3’s output-current monitor and
preferably anywhere else.

It is also convenient and common practice to externally trigger the oscilloscope from the pulse generator’s trigger output. There is nothing wrong with this practice; in fact, it is a recommended approach for ensuring a stable trigger as you move probes between points. This practice does, however, potentially introduce ground loops due to multiple paths between the pulse generator, the circuit, and the oscilloscope. This condition can falsely cause apparent distortion in displayed waveforms. You can avoid this effect by using a trigger isolator at the oscilloscope’s
external-trigger input. This simple coaxial component typically comprises isolated ground and signal paths, which often couple to a pulse transformer to provide a galvanically isolated trigger event.
Commercial examples include the Deerfield Laboratory 185 and the Hewlett-Packard 11356A. Alternatively, you can construct a trigger isolator in a small
BNC-equipped enclosure (Figure A).

Loading comments...

Write a Comment

To comment please Log In