Oscillator meets three requirements
A Gallerani, IRA, CNR, Bologna, Italy
Three common requirements for a clock source are a wide frequency range, a variable duty cycle with independently adjustable TON and TOFF times, and the ability to synchronize with an external signal. The gated oscillator in Figure 1 satisfies all three requirements using just one 74LS123 and a handful of passive components.
To analyze the circuit, first assume that the A input of one-shot IC1A connects to ground. Then, IC1B's positive-going Q output triggers the B input of IC1A, whose negative-going Q output triggers the A input of IC1B. This dc positive feedback ensures that the circuit always self-starts.
The time constant C2X(R2+R4) determines the width of TON, and C1X(R1+R3) determines the width of TOFF. For the 74LS123, the values of the external components at REXT and CEXT essentially define the output pulse width, tW, according to
where k=0.45 for CEXT>1000 pF.
Assuming that RA=R1+R3 and RB=R2+R4, the period and the duty cycle are as follows, respectively:
If C1=C2, then
The circuit oscillates at the frequency
Over the 74LS123's operating range, which is 5 kOhm[less than or equal to]REXT[less than or equal to]200 kOhm and assuming no limits for CEXT, the duty cycle is 100% when RA=5 kOhm and RB=200 kOhm. The duty cycle is 0 when RA=200 kOhm and RB=5 kOhm. And, because TON and TOFF are independent, you can vary the frequency without affecting the duty cycle.
You can easily turn the circuit into a gated oscillator by applying a square-wave gating signal (VC) whose frequency is less than the oscillation frequency (f) to the A input of IC1A. The oscillator output is low when VC is high and is free-running when the gating signal is low. In Figure 1b, the gating signal is 200 Hz, RA=170 kOhm, RB=50 kOhm, and CEXT=2700 pF. With these values, TON[approx. equal to]60 msec, TOFF[approx. equal to]206 msec, and f[approx. equal to]3741 Hz. (DI #2276).
Simple circuit provides digital hysteresis
W Dijkstra, Waalre, The Netherlands
It is sometimes useful to have hysteresis in a digital circuit—for example, in a power circuit under the control of a manual pulse generator, in which mechanical vibrations can produce position errors. The circuit in Figure 1, which consists of a 4585 comparator (IC2), a 4019 switch (IC1), and one-sixth of a 4069 inverter (IC3), provides digital hysteresis. If DI=0, DA=8, and DB=6, then Output 12 and G2 of IC2 assume logic 1, and G1 of IC2 assumes logic 0. Switch IC1 thus connects DA (8) to comparator IC2. When DI=8, Output 12 assumes logic 0 and switch IC1 connects DB (6) to the comparator. If DI is greater than 8, then Output 12 remains at logic 0. If DI becomes less than DB, then Output 12 assumes logic 1. Figure 2 shows a pulse diagram. You can expand the circuit by adding more comparators and switches. (DI #2274).
Generator provides 537 NTSC pattern
Adolfo Mondragon, Philips Components, Juarez, Mexico
If you're involved in the television, CRT, or deflection-yoke business, the circuit in Figure 1 can prove useful for adjusting convergence and purity performance. The circuit generates dots, 537-line- crosshatch, and negative-field (crosshatch- inverted) patterns. You normally use the dots pattern to measure convergence performance using computer-camera equipment, such as the Minolta CC110. The crosshatch also measures convergence but on a visual basis only. The negative field is useful for adjusting and evaluating purity performance.
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Much commercial equipment is available for pattern generation, but the patterns they generate are a mesh of approximately 16 horizontalX16 vertical lines&$151;a lot of lines, when you need only a 535 mesh. In a factory where you must inspect 1000 TVs every day on the production line, a 16X16-line pattern is fatiguing and makes it difficult to concentrate on the critical specification points. All TV, CRT, and deflection-yoke manufacturers specify the convergence- tolerance limits in the intersection of the 535 lines at the edge of the screen. To identify the intersection points at the edges, the pattern in Figure 2 uses "clock" numbers. The generator has the following characteristics:
Its crosshatch pattern uses five vertical and seven horizontal lines.
Its video synchronization uses a noninterlace mode.
In noninterlace mode, its video signal requires no equalization pulses, thereby simplifying the circuit.
Its 262 divider provides a simple way to generate the vertical-synchronization pulses.
Because modern TVs have excellent synchronization circuits, the horizontal oscillator requires no critical, expensive components. In fact, you can set the horizontal oscillator at 15.60 to 15.85 kHz, and the TV synchronizes.
The "brain" of the circuit is the 74HC14 Schmitt-trigger IC. The Schmitt triggers in Figure 1 form both astable oscillators and monostable pulse shapers. The IC provides the 15.75-kHz master horizontal frequency, horizontal lines, and vertical lines. To adjust the circuit, trim PT1 to obtain a frequency that is as close as possible to 15.72 kHz. This frequency, rather than 15.75 kHz, provides a perfect 60-Hz (15,720÷262) vertical pulse. The CD4040 provides the divide-by-262 function. Adjust PT2 to obtain five pulses of 0.5 msec each. These pulses generate the vertical lines in the crosshatch pattern. (DI #2275).
Offline IGBT charges batteries
Christophe Basso, Motorola SPS, Toulouse, France
Power supplies for portable equipment must be light, and they must provide international travelers with a convenient universal input. Although switch-mode power supplies naturally benefit from a universal input, they are more expensive than standard linear supplies, which are based on a transformer. Recent insulated-gate bipolar transistors (IGBTs) offer an inexpensive way to charge batteries from the ac line. The MMG05N60D in Figure 1 sustains as much as 600V and has an avalanche characteristic comparable with that of a MOSFET having the same ratings. However, thanks to the IGBT's small die area, the device costs much less than the high-voltage MOSFET. Moreover, the MMG05N60D's SOT-223 package is pin- and size-compatible with the DPAK industry standard.
Figure 1's offline supply uses a current-mode technique, in which a discrete, double-bipolar thyristor (Q1) forms a latch. You can use off-the-shelf components to generate the 50%-duty-cycle clock, provided that the components can generate a 50-kHz square wave using a low start-up current determined by R3. This design uses a NAND-based Schmitt trigger, extracted from an MC14093. The IGBT's 7-nC gate charge lets you drive the device with simple logic gates. The IGBT receives its bias voltage through R8, a 1-kOhm resistor. This resistor's relatively high value does not disturb the oscillator when the thyristor pulls the gate to ground. When the IGBT turns on, the primary current and the voltage across R4 rise. When the peak current occurs, or when 700 mV appear on C6, Q1 turns on and cuts off the IGBT's conduction. As with any flyback circuit, the energy transfer charges C1 and C2, the output capacitors.
In low-cost structures such as this one, you can easily control the output voltage and current by offsetting the dc level across C6. The offsetting uses an optocoupler driven by a dedicated battery-charger circuit, the MC33341. This IC includes a dual control loop that regulates either the current (sensed by R10) or the voltage (sensed by R11/R12). When the current is below its limit, the MC33341 regulates the output voltage at its nominal value (8V, for example) and allows the output current to increase. When the output current reaches the internal threshold (0.2V/R10), the current loop prompts the optocoupler to transform the supply into a constant-current source. Figure 2 shows the supply's typical transfer function. You can download a Spice model of the MC33341 at http://mot2.indirect.com/models/bin/batmag_ic.html.
The dot-marked terminals of the transformer are wired in an unusual way. You use this atypical wiring technique because connecting a fully discharged battery can present a total short circuit at the output. If you short the output, the flyback auxiliary-winding voltage decreases, but the MC33341's supply disappears, leaving the supply without a current limit. To avoid this scenario, connect the output transformer so that it benefits from both the flyback and the forward voltage. When the output constrains the flyback voltage to a low value, the forward voltage powers the MC33341. The auxiliary winding also benefits from this structure. We tested the supply with a transformer having 6-mH primary inductance and auxiliary- and power-winding ratios of 0.12 and 0.06, respectively. To cancel any turn-on losses, you must prevent the supply from entering the continuous mode. The IGBT typically keeps the current-tail losses at 6 mJ (at IC=0.3A, TJ=125°C, and dVCE/dt=1 kV/µsec). (DI #2297).
MOS transistors form current-mode Schmitt trigger
Tai-Shan Liao, Chun-Ming Chang, and Wen-Yaw Chung, Chung-Yuan, Christian University, Taiwan, China
Schmitt triggers are useful in both analog and digital circuits for reducing sensitivity to noise and disturbances. Current-mode Schmitt triggers are particularly useful in photodetectors, bar-code readers, and optical remote controls. The resistorless current-mode Schmitt trigger in Figure 1 uses six MOS transistors. The circuit uses the output of an inverter and derives its feedback through a variation of threshold current. The circuit uses one inverter pair (QN2, QP4), one current-mirror pair (QP1, QP2), and two load devices (QN1, QP3). The gate of QP3 receives feedback from the output of the inverter. The drain current of QN1 is the lower threshold current (ITL); the drain current of the QN1–QP3 pair is the upper threshold current (ITH). The hysteresis, IH, is ITH–ITL, or (IP3+IN1)–IN1=IP3.
The NMOS and PMOS transistors are components of three CD4007 ICs: IC1, IC2, and IC3. The circuit operates as follows: First, assume that the output of the inverter is in its low state, turning QP3 on. When the phototransistor current (IPD)exceeds the upper threshold (ITH), the output of the inverter switches high, turning off QP3. When IPD falls lower than the lower threshold current (ITL), the inverter switches low, again turning on QP3. Table 1 summarizes the relationships between VDD, ITL, ITH, and IH. The circuit in Figure 1 can operate successfully from a 1.5V supply. (DI #2303).
|TABLE 1—SCHMITT-TRIGGER THRESHOLDS AND HYSTERESIS|
|Power supply||Upper limit||Lower limit||Hysteresis|
|VDD (V)||ITH (mA)||ITL (mA)||IH (mA)|
Eight-channel data-acquisition system features autocalibration
Mark Shill, Burr-Brown Corp, Tucson, AZ
The circuit in Figure 1 is a versatile, eight-channel, differential-input, 16-bit data-acquisition system. A PC controls the circuit via the computer's parallel port. Under control of the PC, you can select one of eight differential voltage inputs, each with a programmable input-voltage range of ±1.25, ±2.5, ±5, or ±10V. In addition, calibration factors stored in the controlling PC's software calibrate each input range for gain and offset errors.
The heart of the system is the ADS7805 16-bit ADC. This ADC can digitize voltages of ±10V as fast as 100 kHz. A unique feature of the ADS7805 allows you to read its 16-bit output as a high byte and a low byte, selected by the Byte pin. When Byte is low, the upper byte of the conversion result is present at data-output pins D15 through D8; when Byte is high, the lower byte appears. This feature simplifies the digital-data interface to the PC's parallel port, in that you need only one 74HC157 quad two-line-to-one-line multiplexer to convert the ADS7805's 16-bit digital data to four 4-bit nybbles. Logic gates IC11 through IC13 create an additional multiplexer channel for the PC to monitor the ADC's end-of-conversion signal BUSY.
A short, active-low pulse applied to the R/C pin starts the conversion. Logic gates IC7 through IC10, with R6 and C1, generate the active-low pulse whenever Strobe pin 1 of the PC's parallel port has a low-to-high transition. For the component values shown, the active-low pulse width is approximately 75 nsec. When the ADC finishes its conversion, which is signified by a low-to-high transition on BUSY, you can read the 16-bit data results as four 4-bit words, under control of the PC's parallel-port output pins 14, 16, and 17 (Table 1).
IC4 is a PGA206 digitally programmable-gain amplifier, which converts the selected differential-input voltage to a single-ended voltage output that connects to the ADS7805 through R5. R5 attenuates the ADC's input signal by approximately 1%, thus allowing the ADC to measure input voltages slightly beyond its normal ±10V input range. The DAC thus has a small overrange capability. The PC can select the PGA206's gain to be 1, 2, 4, or 8V/V, corresponding to input ranges of ±10, ±5, ±2.5, or ±1.25V, respectively. The parallel port's output pins 5 and 6 control the PGA206's address lines A0 and A1 to select the user-defined gain ranges.
IC2, an MPC507 differential, eight-channel multiplexer, selects each of the eight input-voltage channels. Address lines A0, A1, and A2 select the active channel of the MPC507. These address lines connect to the parallel port's output pins 2, 3, and 4, respectively. A logic-high level on the multiplexer's EN pin enables the multiplexer. A logic-low level turns off the mulitplexer's outputs. The gain and offset calibration circuitry comprises IC1 and IC3. IC1 is a precision 5V reference that provides the calibration standard. The 5V, used for PGA gains of 1 and 2V/V, provides corresponding ADC input voltages of 5 and 10V, respectively. In addition, the resistive divider comprising R2 through R4 creates a 1.25V standard that calibrates the 4 and 8V/V gain ranges. Again, the corresponding ADC input voltages are 5 and 10V for the gain ranges of 4 and 8V/V, respectively.
Offset calibration uses ground as the reference for each of the PGA206's gain ranges. IC3 is a differential, four-channel analog multiplexer that switches in each of the calibration-voltage standards to the PGA206's inputs. Because each of the calibration-voltage standards is a known value, you can determine the system's gain and offset errors by digitizing each voltage standard and comparing the result to the theoretical value in the system's software. You can then store the resulting error-correction factors in a program array and use them to correct any measured input voltages in channels one through eight.
Listing 1 shows the Pascal program for the data-acquisition system. For simplicity, the listing shows the data-acquisition part of the program as a unit file, which links to the main program as shown. The program allows you to specify the number of ADS7805 readings to take to average the resulting measurements. A greater number of averages reduces the effects of any noise in either the input signal or the measurement system. The program variable Average sets the number of averages to take. If the input voltage exceeds the selected input-voltage range, the program signifies the overrange condition by writing "Overrange" to the display. (DI #2289).
|TABLE 1—PASCAL LISTING FOR AUTOCALIBRATED DATA-ACQUISITION SYSTEM|
|Parallel-port function||Parallel-port pin||Circuit function|
|Data bit 0||2||Input multiplexer A0|
|Data bit 1||3||Input multiplexer A1|
|Data bit 2||4||Input multiplexer A2|
|Data bit 3||5||PGA206 A0|
|Data bit 4||6||PGA206 A1|
|Data bit 5||7||Input multiplexer enable|
|–Ack||10||Nybble bit 2|
|Busy||11||Nybble bit 3 (MSB)|
|PaperEnd||12||Nybble bit 1|
|Select||13||Nybble bit 0 (LSB)|
|Ground||18 to 25||Ground|
µC provides three-key, five-sequence lock function
William Grill, Riverhead Systems, Littleton, CO
Using MicroChip's 12-C508 eight-pin µC, you can inexpensively implement a digital-sequence lock with debounce and status-indicator features (Figure 1). The design exploits the internal-oscillator, watchdog-timer, and wake-up-on-pin-change features inherent in the µC, and it provides an application that supports key acknowledge and out-of-sequence error detection. The circuit debounces key entries and acknowledges them at Pin 3 with a short LED flash. The µC evaluates the entries against a sequence table that is internally coded to the defined sequence's length. At the end of the sequence, correctly sequenced inputs generate a 2-sec, high-true pulse on Pin 5; a wrong sequence, detected at any time during entry, generates a series of five flashing pulses on Pin 2.
Using the watchdog-timer and wake-up-on-pin-change features to generate timing and initiate processing allows battery-powered operation. The µC also provides an auto-reset function after several seconds between expected entries. The lock-sequence's length and combination are coded characteristics that you establish during programming. Using only 150 bytes of code space, sequences longer than 100 steps are possible. You can also port the code to a larger 16C5x or 16C6x controller and take advantage of the opportunity to include embedded I/O or reporting functions, based on the correctly keyed access processes described here. Click here to download the assembly code for the µC. (DI #2299).